The present disclosure relates generally to multiband transceivers.
The growth of the usage of mobile internet and multimedia services has been explosive in recent years, as witnessed by the spur of user demands such as Web browsing, music download, movie streaming, video teleconferencing, social networking, and broadcast television. As a result, advanced mobile devices have been developed, including smart phones, PDAs, and tablet PCs, to name a few, to provide end users with versatile features and services. These mobile devices are required to support higher data rates promised by 3G WCDMA/HSPA, and even 4G LTE standards, with backward compatibility to the legacy 2G GSM and 2.5G GPRS/EDGE standards. As a recent example, the Apple iPhone 5 and iPhone 4S support quad-band GSM at 850/900/1,800/1,900 MHz, quad-band UMTS/HSDPA/HSUPA at 850/900/1,900/3,100 MHz, and dual band CDMA EV-DO Rev. A at 800/1900 MHz, for a total of six different frequency bands. Furthermore, in the 4G standard, there are more than forty bands all over the world.
While supporting a combination of frequency bands, at the same time, cost and size of mobile devices are also crucial. Discrete power amplifiers have been widely adopted in mobile handsets. For example, one handset can have a quad-band power amplifier module that supports 2G/2.5G/GPRS/EDGE systems and one to five single-band power amplifiers that support 3G WCDMA/HSPA and 4G LTE standards. Although the overall transmission performance is excellent with such multi-module configuration, there is a significant penalty in size and cost.
Converged power amplifiers are proposed to reduce the number of RF signal paths and BOM cost, as well as to simplify routing complexity at the board level. FIG. 1 shows an exemplary block diagram of a transmitter path in a smart phone utilizing converged power amplifiers. This prior art circuitry comprises a transceiver 10, a power amplifier module 12, isolators 18, an antenna switch 14, and a main antenna 16. As shown in FIG. 1, there are only two power amplifiers 22HB and 22LB, one for high band and the other for low band, in the power amplifier module 12 to support multi-mode multi-band (MMMB) outputs. Multi-mode covers multiple standards while multi-band covers multiple bands of outputs. For example, the high-band output, provided by a high-band power amplifier 22HB, supports saturated GMSK mode and linear WCDMA/HSPA mode from 1.71 to 1.98 GHz, while the low-band output, provided by a low-band power amplifier 22LB, covers 2G/2.5G at GSM/EGSM band and 3G from 824 to 915 MHz. A post power amplifier switcher 20 routes the signals properly at each mode and each band of operation.
The transceiver 10 feeds proper RF signals at different frequency bands of operations via RF ports RFOLB or RFOLB to the power amplifier module 12. Typically, in the conventional solution, there are two pre power amplifiers (PPAs), instead of one. The amount of PPAs is proportional to numbers of supported bands or modes. FIG. 2 demonstrates a solution for a transceiver to provide two RF signals paths, capable of being connecting to the high-band and low-band power amplifiers 22HB and 22LB in FIG. 1, respectively. The transceiver includes a PPA 30, an inbound switch circuit 32, transformers 34HB and 34LB, an outbound switch circuit 38, and a pair of high-band and low-band RF ports RFOHB and RFOLB. Based upon band selection, the inbound switch circuit 32 routes the RF signals from the pre power amplifier 30 to the primary winding of either transformer 34HB or 34LB, and the secondary windings of transformers 34HB and 34LB are respectively coupled by the outbound switch circuit 38 to the high-band and low-band RF ports (RFOHB and RFOLO). The transformers, 34HB and 34LB, are independent to each other in view of magnetic coupling. The solution proposed in FIG. 2 is easy to design for band selection and impedance ratio, because optimization of one transformer poses no influences on the other. Nevertheless, a monolithic transformer implemented in an integrated circuit chip always occupies a considerable proportion of the silicon area in a transceiver, and two monolithic transformers, as suggested in FIG. 2, is excessive and costly.
FIGS. 3A and 3B demonstrate two alternatives to the prior art solution suggested in FIG. 2, each employing only one transformer instead of two to save silicon cost. In FIG. 3A, a transformer 36A has one primary winding and two secondary windings, and outbound switch circuit 42 selects and couples one of the two secondary windings based on band selection. A transformer 36B in FIG. 3B has two primary windings and one secondary winding while switch circuits 32 and 40 route RF signal properly depending on either operating at high-band or low-band mode.
While each of FIGS. 3A and 3B has only one transformer, it is inevitable that they come with a penalty in either resonant tuning or loadline impedance optimization. The design difficulties are explained in FIGS. 4A and 4B. FIG. 4A shows a transformer 50 with properly-selected primary and secondary windings (51 and 53) respectively coupled to a PPA 30 and a load resistor 52. The load resistor 52 denotes the characteristic impedance RL of traces of a selected RF port. An equivalent circuit of FIG. 4A is shown in FIG. 4B, where VG represents the driven output voltage swing by PPA 30, RG represents the equivalent output impedance of the PPA 30, CP represents the capacitance of the tunable capacitor 55, rP represents the input parasitic series resistance of the primary winding 51, km represents the coupling coefficient of the transformer 50, LP represents the inductance of the primary winding 51, (1−km2)×LP represents the leakage inductance of the primary winding 51, km2×LP represents the mutual-coupling inductance of the transformer 50, n represents the turn ratio between the primary and secondary windings, and RL/(n/km)2 represents the effective load impedance transferred by transformer 50 from the secondary side to the primary side. The primary winding 51 and the tunable capacitor 55 of FIG. 4A form a LC tank, whose resonant frequency determines the center frequency of a selected band. Based on the solution shown in FIG. 3A, where LP is a constant, if the high-band frequency is 3 times higher than that of the low band, the capacitance CP of the tunable capacitor 55 needs to be tuned by 9 times to switch between high and low bands. To cover such a large capacitance tuning ratio, a large silicon area with a high die cost is required for the tunable capacitor in FIG. 3A, making the solution in FIG. 3A non-preferable. Furthermore, the quality factor of the capacitor would be low resulting in high loss and degraded performance. FIG. 3B avoids the disadvantage of FIG. 3A by using two primary windings with inductances different to each other. Accordingly, the required capacitance tuning ranges of the two capacitors in FIG. 3B can be significantly smaller. For example, the inductance LP of the primary winding 51 in FIG. 4B becomes 9 times larger and CP is kept approximately the same, to form a LC tank for a high band 3 times higher. It is worth noting that LS, the inductance on the secondary side, is always the same for both high-band and low-band operations because of the common secondary winding in FIG. 3B. It can be concluded from FIG. 4B that in this scenario, the turn ratio n triples, the loadline impedance RL/(n/km)2 decreases, and the voltage drop VCOUPLE across the loadline impedance RL/(n/km)2 changes. As the voltage drop VCOUPLE correlates to the total output power delivered by the transformer, power added efficiency (PAE) for the solution in FIG. 3B cannot be simultaneously optimized for both high and low bands.